Switching Power Converters have for many years served as a viable means for electrical energy conversion. Unfortunately, although the semiconductor devices used in these converters are operated in a manner similar to that of switches, undesirable energy dissipation internal to these conventional devices nevertheless occurs during turn-on and turn-off transitions. Such losses are due to the simultaneous existence of voltage across and current through the semiconductor devices during commutation. Because these losses occur at each switch transition, high frequency operation correspondingly yields low power conversion efficiencies.
Since higher switching frequencies generally result in smaller reactive components and improved dynamic performance, mechanisms for minimizing switching losses have long been sought after. For example, in conventional Pulse-Width-Modulated (PWM) switch-mode converters, energy recovery snubbers have been used to "soften" the switching of semiconductor devices. A technique known as "soft-switching" has been implemented in switching power converters. This conventional technique seeks to eliminate switching losses by altering the switching conditions in such a way that the switch current or switch voltage is zero at the time of commutation.
In this way "Zero-Current-Switching" (ZCS) or "Zero-Voltage-Switching" (ZVS) respectively, is attempted. To implement this switching mechanism, an L-C network is added around the switch so that the switch current or switch voltage may be kept at a constant zero value during switch commutation. Conventional switch-mode converters using this type of soft-switch are known as Quasi-Resonant Converters (QRC).
ZCS may be attempted when an inductor is placed in series with the semiconductor switch (FIG. 1). Since the energy stored in an inductor cannot change instantaneously, neither can the current through it change instantaneously. If energy resonates between the inductor and the capacitor when the switch is on, then the switch may be opened losslessly (in theory) at a time when the inductor has dumped all of its energy to the capacitor. Once the switch is open, the inductor current remains zero, and the switch can turn on with zero-current through it.
ZVS may be attempted when a capacitor is placed in parallel with the semiconductor switch (FIGS. 2 and 3). Since energy stored in the capacitor cannot change instantaneously, neither can the voltage across it change instantaneously. If energy resonates between the capacitor and the inductor when the switch is off, then the switch may be closed losslessly (in theory)at a time when the capacitor has dumped all of its energy to the inductor. Once the switch is closed, the capacitor is shorted and its voltage remains zero, thus allowing zero-voltage-turn-off for the switch.
Both ZVS and ZCS conventional techniques seek to decrease switching losses and attempt to permit high efficiency operation at higher switching frequencies. However, only ZVS is effective in reducing switching losses at high frequencies, because some loss occurs during turn-on of a ZCS switch. Parasitic capacitance across the semiconductor switch stores energy while the switch is off, and releases it internally when the switch is turned on. For this reason, high frequency operation of such conventional converters, even with the attendant switching losses, is possible only with ZVS converters.
In practice, ZCS techniques have often been used in place of ZVS techniques even though switching losses are not altogether eliminated (e.g. ZCS Boost QRC of FIG. 7), because with ZVS converters, the large resonant voltage of the resonant capacitor is imposed across the active switch. In some half and full-bridge converter topologies, this resonant voltage is limited by the clamping action of the input voltage source. In known single-ended converter topologies, such as the ZVS Boost QRC of FIG. 8, the voltage is unrestrained and may peak at a value equal to ten times or more the input or output voltage. This peak resonant voltage is also a strong function of the output load resistance or current. Therefor, at high voltage and/or high power levels, the voltage stresses impressed upon the active switch are intolerable, thus making ZVS implementation in single-ended converters impractical.
Several techniques have attempted to reduce these high voltage levels in the hopes of making ZVS a viable technique for high voltage and/or high power applications. One such technique known as ZV Multi-Resonant Switching (ZVS-MR) reduces voltage stresses by adding a second resonant capacitor across the rectifying diode(s) of the power converter. Two resonant capacitors are provided, one across the active switch(es), and the other across the rectifier diode(s) which share the energy resonating from the resonant inductor. In this way, the peak voltage across the active switch is reduced since the high voltage is divided between the two resonant capacitors. A typical conventional ZVS-MR Boost Converter is shown in FIG. 9. Unfortunately, this technique does not eliminate these stresses, particularly so for off-line applications where input voltages may be as high as several hundred volts.
Another technique incorporating the above mentioned multi-resonant technique has also been used to decrease the voltage stresses on the active switch. This technique utilizes the above mentioned two capacitor multi-resonant circuit to include a voltage clamping mechanism to limit the active switch voltage. This technique implemented in a Boost converter is shown in FIG. 10. The voltage clamp includes a bulk capacitor and an auxiliary switch. Since the bulk capacitor is large relative to the resonant capacitors, the voltage across it can be approximately constant over a switching cycle. When the voltage on the active switch is equal to that of the bulk capacitor, the auxiliary switch turns on with ZVS and energy flowing from the resonant inductor is routed from the resonant capacitor to the bulk capacitor.
The auxiliary switch turns back off once the amount of charge in the bulk capacitor has flowed back out. In this way no net charge accumulates on the bulk capacitor from one cycle to the next and its voltage remains essentially constant. This technique lowers peak voltage stresses, but circuit complexity is increased, and reliability is correspondingly decreased since failure of the active voltage clamping circuit would cause voltage breakdown in the main active switch and consequent failure of the power supply itself.
One type of conventional converter which has successfully addressed low voltage stress operation for all active and passive semiconductor devices is known as the Quasi-Square Wave (QSW) Converter (FIG. 11). This converter modifies the switch-mode single-ended converters by placing a resonant capacitor across the active switch and/or the passive switch (diode); the filtering inductor is replaced by a small resonant inductor. A diode is added in parallel with the active switch, and a second active switch is sometimes added across the rectifying diode (FIG. 11). In doing so, all semiconductor switches operate with near ZVS, and their peak voltage is passively limited (by the diodes in the circuit) to whatever voltage sources and sinks are present in the circuit. For example, in a Boost converter, the input filter inductor is replaced by a resonant inductor, and the voltage stress on the switches is equal to the output voltage. Unfortunately, rms currents in the resonant inductor are unacceptably high, and an essentially constant resonant inductor current is impossible. In effect, a QSW Boost converter is no longer driven by an effective current source, but rather a voltage source.
In a quasi-square wave boost converter, the voltage stress of the active switch and passive switch is limited to the output voltage since these switches along with the output filter capacitor form a closed loop. In other words, the sum of the two switch voltages equals the output voltage. The two diodes in the circuit will passively turn-on when the voltage on either switch reaches the output voltage. To conserve the original operation of the Boost converter, the input inductor must be restored to a filter inductor (so that input current may be continuous and nearly constant if desired,) and the resonant inductor must be moved to a new location in the circuit.
For ZVS operation of the switches a resonant inductor is needed to remove the charge stored within the parasitic capacitances of each switch. By adding an auxiliary switch and diode, a zero-voltage-transition (ZVT) converter circuit of FIG. 12 is known. The ZVT converter achieves ZVS operation for the main power switch S and power rectifier D.sub.R, however, the auxiliary switch and diode operate with ZCS. Although the main power flow is not directed through these devices, nevertheless losses can be unacceptably significant since the voltage across the devices may be as high as 400 Volts in universal input off-line applications. This lossy switching results in the inability to operate at very high frequencies.
Therefor, single-ended switching power converters possessing exclusively ZVS operated semiconductor devices, along with low voltage stresses have not been forthcoming. A ZVS-MR converter with an auxiliary active voltage clamping mechanism is not reliable since the clamping is not passive but active, and added complexity is also required for controlling its auxiliary clamping switch. Quasi-square wave converters possess an inherent passive voltage clamping mechanism, however the basic operation of these converters have been altered from that of their switch-mode counterparts. Input or output filter inductors have been replaced by small resonant inductors causing high rms currents. By relocating the resonant inductor and adding an auxiliary active switch and diode, the ZVT converter provides desired ZVS switching for the power switch and power rectifier, but operates the auxiliary active switch and diode with ZCS. This yields significant turn-on losses at high frequencies.
Low loss switching techniques are desirable for improving converter efficiencies, however implementing these conventional schemes have been problematical. For example, ZVS techniques usually result in high voltage stresses in single-ended converters. Another problem concerns the means by which output voltage or current regulation is accomplished. In conventional Switch-Mode converters, regulation is achieved by varying the switch duty cycle while maintaining the switching frequency constant. In conventional soft-switching converters, the control law for regulating output voltage or current has inadvertently changed from a constant frequency to a variable frequency control scheme. This is an undesirable control method, making line filter designs more difficult as well as adversely affecting electromagnetic interference. Therefore, methods for maintaining constant frequency control, along with soft-switching, have been sought after.
There is a fundamental reason why conventional quasi-resonant converters must operate with a variable frequency control law: In the case of ZCS, the switch may turn on at any time with zero current turn-on, but the turn-off of the switch is determined by the resonant inductor current. In the case of a bidirectional switch, the turn-off must occur when the resonant inductor causes current to flow through the switch parallel diode. As a result, the on-time of the switch is determined by the resonant L-C circuit as well as the operating point (i.e. input and output voltage and current). If the effective duty cycle is to be changed, it must occur by changing the switching period rather than the on-time. In the case of ZVS, the switch may turn off at any time with zero-voltage turn-off, but the turn-on of the switch is determined by the resonant capacitor voltage. For a uni-polar switch, the turn-on must occur when the resonant capacitor causes the voltage across the switch to go to zero and thus turn on the parallel diode. Therefor, the off-time of the switch is not controllable, thereby forcing the switching period to become the controlled variable for achieving voltage regulation.
By implementing such soft-switching, freedom in control is lost. In conventional switch-mode converters, both the on-time and the off-time of the switch is controllable, but with soft-switching, either the on-time (ZCS) or the off-time (ZVS) is no longer arbitrarily controlled. To achieve soft-switching along with constant frequency control, freedom of control must be re-established.
Many topologies are known which seek to implement soft-switching and restore constant-frequency control. These conventional implementations use at least two active switching devices operating with either ZCS or ZVS. One such family of constant frequency controlled converters have been referred to as Extended Period Quasi-Resonant Converters, because the resonant cycle between L.sub.r and C.sub.r which is present in all QRC's, is temporarily interrupted by the opening or closing of an auxiliary switch. The equivalent on-time (for ZCS) or off-time (for ZVS) of the switch is determined by the L-C components as well as the input and output voltages and currents. This on-time or off-time is not a controllable parameter, and as a result the output is controlled only by varying the switching frequency. By interrupting the resonant cycle, the equivalent on-time or off-time may be extended for an arbitrary amount of time. In this way, freedom to control both on-time and off-time is restored, and constant frequency operation is made possible.
Extended period quasi-resonant converters (QRC's) are either: parallel mode, or series mode. In parallel mode extended period QRC's, the auxiliary switch which interrupts the resonant cycle is in parallel with one of the resonant components and commutes with ZVS. Two such Boost converters are depicted in FIG. 13 and 14. In series mode extended period QRC's, the auxiliary switch is in series with one of the resonant components, and commutes with ZCS (FIG. 15). Extended period QRC's may be implemented in both ZCS and ZVS conventional QRC's. As a result, one switch might operate with ZVS, while the other operates with ZCS, or vice versa, or both switches may be operated with ZCS or ZVS.
Extended period quasi-resonant converters exhibit constant frequency control while maintaining soft-switching on all switches. However, limitations in load and line regulation are similar to those of conventional QRC's. It is well established that QRC's operated in a full-wave mode exhibit significantly improved load regulation over QRC's operated in a half-wave mode. Similarly, extended period quasi-resonant converters operating in a full-wave mode exhibit far superior load regulation over similar converters operated in a half-wave mode. In extended mode QRC's, several problems make ZVS operation impractical.
First, although full-wave mode operation is desirable for good load regulation, it is undesirable for true ZVS: parasitic capacitance across the active power switch retains charge trapped when the series diode turns off which yield unacceptable turn-on loss similar to the loss in ZCS. Secondly, high voltage stresses present across the switches make such converters impractical. Therefor, although conventional extended mode QRC's can provide constant frequency operation, practical ZVS implementations which are applicable in high power or high voltage systems are not forthcoming. The ZCS extended mode QRC's however, do provide advantages in terms of voltage stresses and load regulation. When operated in a full-wave mode, load and line regulation is comparable to their Switch-Mode counterparts.
Unfortunately, at higher power levels, dv/dt across the switch becomes very large causing unacceptable voltage overshoot, as well as high frequency oscillation between the resonant inductor and parasitic switch capacitance. Passive or active snubbers may alleviate this problem, only to lead to either loss of efficiency or added complexity, or both.
Another conventional method of producing soft-switching constant frequency controlled single-ended converters modifies a variable frequency controlled ZVS-MR converter as shown in U.S. Pat. Nos. 4,841,220, 4,857,822, and 4,860,184. By adding an active switch across the passive rectifier diode, power flow to the load is controlled via constant frequency control. Such a Boost converter is shown in FIG. 16. Two control methods are known. The first method uses a primary switch to maintain a constant energy storage level in the resonant inductor at the beginning of each switching cycle. The active rectifier switch controls the portion of time during a switching cycle that the L-C tank is connected to the load and supplying power to it. This technique can allow very high frequency operation. Unfortunately, high voltage stresses make these conventional converters impractical for high voltage and/or high power applications.
The second method of control, as shown in U.S. Pat. No. 4,931,716, utilizes an active rectifier switch to control the direction of power flow between the L-C tank and the load. By leaving this switch on longer, more power returns to the tank after having flowed to the load. By leaving the switch on a shorter period of time, less power returns and thus a higher output voltage is obtained for a given output load. This converter and its control technique suffer from both high voltage stresses, and higher rms currents, since bi-directional power to and from the load occur.
As mentioned previously, Quasi-Square Wave converters are known that address the detrimental high voltage stress of single-ended ZVS converters. These converters replace the filtering inductor of the converter with a small resonant inductor. In addition to providing low switch voltage stresses, these converters can approach constant frequency control. When an active switch is added across the rectifying diode, power flow from the source to the resonant inductor at the beginning of the switching cycle may be controlled arbitrarily, and power flow between the resonant inductor and the output load may also be controlled arbitrarily.
However, one problem with this approach is that all currents in the converter have large rms values since the energy storage level in the resonant inductor must cross zero during each switching cycle. Although these known converters physically resemble their switch-mode counterparts, they nevertheless operate very differently. The Boost converter, for instance, may no longer be designed to operate with a near constant input current. Naturally, there exists a non-zero average DC current, however, high harmonics and high rms values make high power factor and high efficiency difficult at medium to high power levels.
A Switch Module is known which provides ZCS turn-on and ZVS turn-off as the active switch of an switch-mode converter. This switch module includes two active (semiconductor) switches, four diodes (only two if two MOSFETS are used as active switches,) and a resonant inductor and resonant capacitor. In a Boost converter implementation, this circuit appears as in FIG. 17. The four diodes clamp the switches' voltage to whatever voltage sources are in the circuit (e.g. V.sub.in for the Buck converter), and the two active switches operate off of the same driving waveform. In principle, this converter may be used at high voltage and high power levels, however ZCS turn-on losses and conduction losses are severe at higher power levels because the input current has to flow through both active switches as well as the resonant inductor during the converter's equivalent on-state.
Several Zero-Voltage-Transition (ZVT) Switching techniques have been used in single-ended switch-mode converters (Boost converter circuits shown in FIGS. 12 and 18) quasi-square wave converters are a type of ZVT converter, since the rate of change of voltage across the active switch and rectifier are equal because the voltage across these sums to a constant voltage. For example, in a Boost converter, the voltage across the active switch, plus the voltage across the rectifier, equals the output voltage. In conventional ZVS QRC's, a separate resonant capacitor is required across the rectifier to produce the ZVS MR QRC's to eliminate unwanted oscillations between the rectifier parasitic capacitance and the resonant inductor.
One should note that the basic principle in ZVT converters is to attempt to provide a resonant inductor which can store sufficient energy such that current flowing through it can--at the desired time--remove the charge stored in the parasitic capacitance of the active switch prior to its commutation to the on state.
Unfortunately, known methods require the addition of an auxiliary switch which operates with ZCS. This auxiliary switch is turned on with ZCS to initiate the removal of charge stored in the parasitic capacitances. One of ordinary skill hopes that switching losses in the auxiliary switch will not be severe since a relatively small portion of average energy flows through it over a switching period. At medium frequencies, the converter operates efficiently, but at frequencies near 1 MHz, turn-on loss are unacceptably significant (25 Watt for V.sub.ds =400 V and C.sub.ds =300 pF.sub.). As mentioned previously, unwanted oscillations are always present with known ZCS techniques. These unwanted oscillations worsen at higher switching frequencies and further deteriorate conventional converter performance.